外文翻譯---單級(jí)單刀功率因數(shù)校正器 ac dc變換器

1、<p><b>　　電子與信息工程學(xué)院</b></p><p>　　本科畢業(yè)論文(設(shè)計(jì))</p><p>　　外 文 文 獻(xiàn) 翻 譯</p><p>　　譯文題目：single-stage single-switch powerfacter correction AC/DC converter</p><p>

2、　　學(xué)生姓名： </p><p>　　專 業(yè)： </p><p>　　指導(dǎo)教師： </p><p>　　2010年 3 月</p><p><b>　　原文:</b></p><p>　　Single

3、-stage single-switch power factor correctionAC/DC converter</p><p><b>　　Abstract</b></p><p>　　A single-stage single-switch power factor correction AC/DC converter is proposed in whi

4、ch the power factor correction (PFC) inductor is connedted to a tap on the primary winding ofthe DC/DC flyback converter stage; there is direct energy transfer of a part of the input energy tothe output and theDC-bus vol

5、tage feedback. The additional discharge path in the PFC inductorand DC-bus voltage feedback effectively suppresses the DC-bus voltage and increases the overallefficiency. Experimental results </p><p>　　1 Int

6、roduction</p><p>　　Many PFC AC/DC converters have been presented inrecent years. PFC techniques can be divided into twocategories: a single-stage approach and a two-stageapproach. The two-stage approach is t

7、he most commonlyused approach. Conventional two-stage PFC convertersinvolve the use of cascading two power-processing stages,responsible for power factor correction and output voltageregualtion. A PFC converter is adopte

8、d at the front-end toforce the line current tracking the line voltage and anotherconventiona</p><p>　　2 Analysis of proposed converter</p><p>　　Figure 1 shows the equivalent circuits of the prop

9、osedconverter. The secondary winding Nb is added in the PFC boost inductor. The first transformer T1 in DC/DC part can be operated either in CCM or DCM depending on the load conditions as in the conventional flyback conv

10、erter. For simplification, DC/DC part is assumed to operate in CCM for entire line period. The second transformer T2 is operated in DCM. According to the operation of T2, When the DC-bus voltage feedback value V1(=np2Vab

11、, np2=N2/N</p><p>　　Fig. 1 Equivalent circuit</p><p>　　2.1 Region A operation</p><p>　　In region A the input voltage vi is lower than the DC-bus voltage feedback value Vl=(N2/Np)Vd.

12、 Only the DC/DC part operates. It delivers power from the DC-bus capacitor Cd to the load RL through T1. At to the switch S is turned on. Since the DC-bus voltage Vd is applied across the magnetising inductance Lm1, the

13、magnetising current iLm1 increases linearly from its lowpeak value ImA,L as follows:</p><p>　　At t1 the switch S is turned off and the output diode D1 is on. Since -npVo is applied across the magnetisingindu

14、ctance Lm1, the current decreases linearly from its highpeak value ImA,H as follows:</p><p>　　The diode current iD1 is given by</p><p>　　From (3) and (4) the voltage gain is determined as follow

15、s:</p><p>　　Since the DC/DC part operates in CCM, the duty cycle D does not change with the load variation. From (6) the turn ration np can be determined by</p><p>　　The duty cycle D can be obta

16、ined as</p><p>　　The duty cycle D is also effective in regions B and C. The power P1 delivered by the transformer T1 is determined as</p><p>　　Since the power P1 delivered by the transformer T1

17、should be equal to the output power Po (=Vo2 /RL), the following relation is obtained:</p><p>　　Consequently the DC-bus capacitor provides the whole power to the load. From (3), (6) and (10), the high and lo

18、w peak values of iLm1 are determined by</p><p>　　2.2 Region B operation</p><p>　　When the voltage vi is higher than V1 and lower than V2, the converter operates in region B. PFC cell operates as

19、 DCM flyback converter. At t0 the switch S is turned on. Since the DC-bus voltage Vd is applied across the magnetising inductance Lm1, the magnetising current iLm1 increases linearly from its low peak value ImB,Las follo

20、ws:</p><p>　　The magnetising current iLm2 increases linearly from zero as follows:</p><p>　　Sincethe DC-bus capacitor current iCd, the current i2, and the switch current is are given by</p>

21、;<p>　　From (15) it can be seen that the magnetising current iLm1 is supplied by the magnetising current iLm2 and the discharging current iCd of the DC-bus capacitor Cd. From (17) the switch current is is also com

22、posed of the two components. The condution loss can be reduced by selecting smaller np1. At t1 the switch S is turned off and the output diodes D1 and D2 are on. Since npVo is applied across the magnetising inductance Lm

23、1, the current iLm1decreases linearly from its high peak value ImB,H asfol</p><p>　　Since the diode D2 is on, _naVo is applied across the magnetising inductance Lm2 and the current iLm2 decreases linearly fr

24、om its peak value Im,p as follows:</p><p>　　The peak value Im,p is given by</p><p><b>　　From (20)</b></p><p>　　At t2 the current iLm2 arrives at zero and the diode D2 is

25、turned off. The power P1 delivered by the transformer T1 is given by</p><p>　　The power P2 delivered by the transformer T2 is determined by</p><p>　　Since the sum of the power P1 delivered by th

26、e transformer T1 and the power P2 delivered by the transformer T2 should be equal to the output power Po, the following relation is</p><p><b>　　obtained:</b></p><p>　　From (6), (13)

27、and (24), the high and low peak values of iLm1 are determined by</p><p>　　2.3 Region C operation</p><p>　　When the voltage vi is higher than V2 the converter operates in region C. The PFC cell

28、operates as a DCM boost converter and the diode D2 is off in this region. At t0 the switch S is turned on, iLm1 increases linearly as</p><p><b>　　follows:</b></p><p>　　iLm2 increases

29、 linearly as in region B. Equations (9)–(11) are also effective in this region. The magnetising current iLm1 is supplied by the magnetising current iLm2 and the discharging</p><p>　　current iCd of the DC-bus

30、 capacitor Cd. When the current iLm2 is high enough that np2iLm2 exceeds iLm1, the DC-bus capacitor can be in charging mode during the ontime interval of S. This charging mode may occur near the line peak voltage as foll

31、ows:</p><p>　　At t1 the switch S is turned off and the output diodes D1.Since _npVo is applied across the magnetising inductance Lm1, the current iLm1 decreases linearly from its high peak value ImC,H as fol

32、lows:</p><p>　　Since the diode Db is on, the voltage across the magnetising inductance Lm2 is _Vd_n1Vo+vi and the current iLm2 decreases linearly from its peak value Im,p as follows:</p><p>　　Th

33、e peak value Im,p is given by</p><p><b>　　From (31)</b></p><p>　　Sincethe current i1, the DC-bus capacitor current iCd, and the diode current iD1 are determine as follows:</p>

34、<p>　　The current iD1 is composed of two components which are from Lm1 and Lm2. Therefore, there is direct power transfer from the line input to the load during the switch off-time. As a result the overall efficien

35、cy can be improved. At t2, the current iLm2 arrives at zero and the diode Db is turned off. The current iD1 is npiLm1. Since the output power should be equal to the sum of the power from Lm1 and the power from Lm2, the f

36、ollowing relation is</p><p><b>　　obtained:</b></p><p>　　From (6), (13) and (24), the high and low peak values of iLm1 are determined by</p><p>　　3 Concluding remarks<

37、/p><p>　　A single-stage single-switch power factor correction AC/DCconverter has been proposed. Experimental results for a60Wconverter at a constant switching frequency of 70 kHz have been given to show the per

38、formance of the proposed converter. Experimental results have shown that the voltageacross the DC-bus capacitor can be held below 404V even though the converter operates in a wide range of inputvoltages (90B265 V). As a

39、result, commercially availableelectrolytic capacitors, can be used. The CCM DC/</p><p><b>　　譯文:</b></p><p>　　單級(jí)單刀功率因數(shù)校正器 AC / DC變換器</p><p><b>　　摘要</b></p&

40、gt;<p>　　單級(jí)單刀開(kāi)關(guān)AC / DC功率因數(shù)校正器（PFC）是電感連接到初級(jí)繞組的直流/直流轉(zhuǎn)換器；輸入能量的一部分直接轉(zhuǎn)移成輸出能量和反饋給DC總線。另外在PFC電感放電和DC總線電壓反饋過(guò)程中，有效地抑制了直流母線電壓，提高了整體效率，在一個(gè)功率60W、恒定開(kāi)關(guān)頻率70千赫轉(zhuǎn)換器的實(shí)驗(yàn)中的性能結(jié)果表明，電壓在直流總線下電容可以兼容，即使低于405V，轉(zhuǎn)器工作在帶寬輸入電壓（90-265 VAC）的條件下，測(cè)量輸

41、入電諧頻各項(xiàng)指標(biāo)滿足國(guó)際電工委員會(huì)要求。</p><p><b>　　1簡(jiǎn)介</b></p><p>　　近年來(lái)，許多功率因數(shù)校正AC / DC轉(zhuǎn)換器相繼問(wèn)世。 PFC的技術(shù)可以分為兩個(gè)類別：?jiǎn)渭?jí)方法和雙級(jí)方法。這兩個(gè)不同層次的方法是最常使用的方法。傳統(tǒng)的雙級(jí)PFC變換器涉及兩個(gè)級(jí)聯(lián)在電路中使用，負(fù)責(zé)功率因數(shù)校正和輸出額定電壓 。PFC轉(zhuǎn)換器是采用在前端迫使線電流跟蹤

42、線電壓，一種傳統(tǒng)的直流/直流轉(zhuǎn)換器串聯(lián)后的PFC可以獲得現(xiàn)階段所需的穩(wěn)定的額定輸出電壓。這種方法可以取得很好的成效，如提高功率因數(shù)，降低電壓沖擊。然而，由于兩種電源的工作轉(zhuǎn)換使用效率降低，增加了PFC階段額外的組件和復(fù)雜性,因此整體成本增加。這兩個(gè)階段的方法具有低功率強(qiáng)度，控制復(fù)雜，成本高的缺點(diǎn)。 為了減少整體規(guī)模和成本，在一些相關(guān)著作中對(duì)很多單級(jí)PFC的轉(zhuǎn)換器作了詳細(xì)闡述。其主要思想是，PFC和DC / DC有共同的PF

43、C開(kāi)關(guān)，它的開(kāi)關(guān)及其控制器可以被閑置。由于單級(jí)方法簡(jiǎn)單特別是在低成本和低功耗地方得到了應(yīng)用。然而，它仍然有幾個(gè)缺點(diǎn) 比如在電源開(kāi)關(guān)和高直流母線電壓沖擊下有較高的電流脈動(dòng)。其主要缺點(diǎn)是對(duì)直流總線電容有高電壓損傷。許多PFC和C / DC轉(zhuǎn)換器受到低負(fù)載和高直流限制，高直流總線高電壓組合方法具有高成</p><p><b>　　2轉(zhuǎn)換器的分析建議</b></p><p>

44、　　圖1顯示了轉(zhuǎn)換器的等效電路圖。二次繞組中添加Nb ，是PFC 的升壓電感。一次側(cè)變壓器T1在DC / DC部分無(wú)論是CCM或DCM，可以根據(jù)負(fù)載在常規(guī)條件下反饋給轉(zhuǎn)換器。為了簡(jiǎn)化，直流/直流部分用于CCM總線路的操作中。變壓器T2的在DCM中。根據(jù)T2的運(yùn)作，有三個(gè)操作，如圖3所示，當(dāng)直流總線V1的電壓反饋值 (=Np2Vd, Np2=N2/Np)超過(guò)輸入電壓的整流線的電壓，該轉(zhuǎn)換器就會(huì)在區(qū)域A運(yùn)行。在A區(qū)只有T1是正在運(yùn)行，Db是

45、在一段時(shí)間s上反向偏置。Vi高于V1和低于V2的時(shí)候，此時(shí)轉(zhuǎn)換操作進(jìn)入B區(qū)。在這個(gè)區(qū)域，T1和T2工作起來(lái)就像一個(gè)回掃轉(zhuǎn)換器; V2是由Vd+(n1-na)Vo決定。其中n1=N1/Ns 和na=Na/Nb。當(dāng)Vi>V2時(shí)該轉(zhuǎn)換器操作進(jìn)入C區(qū)。在這個(gè)區(qū)域，T1就像一個(gè)回掃變壓器在工作并且T2就像一個(gè)升壓電感。由于Vi電壓是正弦曲線最大值的四分之一，是介于tx和tY的，如下所示 ： </p><p>　　圖1

46、轉(zhuǎn)換器的等效電路圖 </p><p><b>　　2.1 A區(qū)操作</b></p><p>　　在區(qū)域A，輸入電壓Vi低于直流總線V1的電壓反饋值，Vl=(N2/Np)Vd.，只有直流/直流部分運(yùn)作。它提供的直流電源通過(guò)T1使總線電容從Cd的負(fù)荷傳給RL。在t0時(shí)刻開(kāi)關(guān)S是打開(kāi)的。由于直流母線電壓Vd適用于整個(gè)磁化電感Lm1中，磁化電流iLm1從最低的IMA，L的峰值

47、線性增加如下：</p><p>　　磁化電感L，從最在T1時(shí)刻開(kāi)關(guān)S是關(guān)閉的，輸出二極管D1處于工作狀態(tài)。由于npVo是適用于整個(gè)高的IMA，H的峰值線性減少如下：</p><p>　　二極管電流ID1是：</p><p>　　在（3）及（4）中電壓V0 如下： </p><p&

48、gt;　　由于直流/直流部分在CCM中運(yùn)作，環(huán)流D不會(huì)改變負(fù)載的變化。從（6）看出np可確定為：</p><p><b>　　環(huán)流D的表達(dá)式為：</b></p><p>　　環(huán)流D 在B區(qū)和C區(qū)也是有效的通過(guò)T1得到的P1的表達(dá)式如下所示：</p><p>　　功率P1等于輸出功率P0即：</p><p>　　因此直流總

49、線電容向負(fù)載提供了全部功率。從（3），（6）及（10），最大值ImA,H和最I(lǐng)mA,L小值取決于</p><p>　　2.2 B區(qū)的運(yùn)作</p><p>　　當(dāng)電壓vi高于V1 或低于V2的時(shí)候，該轉(zhuǎn)換器就在B區(qū)運(yùn)作,此時(shí)PFC單元作為DCM回掃式轉(zhuǎn)換器。在T0時(shí)刻開(kāi)關(guān)S是打開(kāi)的。由于直流母線電壓VD的應(yīng)用整個(gè)磁化電感Lm1被磁化,iLm1從最小值ImB,L線性增加，關(guān)系式如下：<

50、/p><p>　　該磁化電流iLm2從零線性增加，即：</p><p>　　由于N1i1 + N2i2=0的i2－i1=iLm2，直流總線電容電流ICD，，電流I2和開(kāi)關(guān)電流is關(guān)系如下</p><p>　　從（15）可以看出，磁化電流iLm1和磁化電流iLm2 是由直流總線電容ICD的放電提供的。（17）開(kāi)關(guān)電流is也是其中的兩個(gè)組件。通過(guò)選擇np1可以使該傳導(dǎo)的損失

51、減少。在T1時(shí)刻開(kāi)關(guān)S是關(guān)閉的，輸出二極管D1和D2處于工作狀態(tài)。相關(guān)表達(dá)式如下：</p><p>　　由于二極管D2處于工作狀態(tài)，-nav0是從磁化電感Lm2和當(dāng)前電流iLm2的峰值imp線性下降，即：</p><p><b>　　峰值imp為：</b></p><p><b>　　從（20）</b></p>

52、<p>　　在T2時(shí)刻電流iLm2到達(dá)零，二極管D2斷開(kāi)。由T1得到的P1為</p><p>　　由T2 得到的P2</p><p>　　由T1得到P1 和T2得到P2 共同控制輸出電壓，關(guān)系式為： </p><p>　　從（6），（13）和（24），最高和最低峰值iLm1取決于</p><p><b>　　2.3

53、C區(qū)運(yùn)作</b></p><p>　　當(dāng)電壓vi>V2時(shí)，轉(zhuǎn)換器在C區(qū)運(yùn)作，在這個(gè)區(qū)，PFC單元就像DCM升壓轉(zhuǎn)換器同時(shí)二極管D2是斷開(kāi)的。在T0時(shí)刻，開(kāi)關(guān)S是打開(kāi)的，iLm1是線性增加的即：</p><p>　　iLm2在區(qū)域B線性增加。此時(shí)可以運(yùn)用上述的方程組（9） - （11）。磁化電流iLm1是由磁化電流和放電iLm2直流總線電容Cd所提供的。當(dāng)電流iLm2是足夠

54、高并且np2iLm2超過(guò)iLm1時(shí)，直流總線電容在充電過(guò)程中會(huì)出現(xiàn)一個(gè)瞬時(shí)電流。這種充電模式會(huì)產(chǎn)生線性峰值電壓，如下：</p><p>　　在t1時(shí)刻，開(kāi)關(guān)S和輸出二極管D1是斷開(kāi)的。由于-npVo是適用于整個(gè)磁化電感Lm1，電流iLm1線從峰值線性減少</p><p>　　由于二極管DB是連通的，兩端電壓的磁化電感Lm2是-Vd-n1Vo+vi ，電流iLm2從峰值im，p線性減少<

55、;/p><p>　　峰值im，p由下式確定：</p><p><b>　　由 (31)：</b></p><p>　　由于 電流i1，直流總線電容電流iCd和二極管電流iD1的決定式如下</p><p>　　電流ID1來(lái)自于電感 Lm1和Lm2兩個(gè)部分。因此，功率直接從輸入線路到負(fù)載上轉(zhuǎn)換，因此總體效率被提高。在t2時(shí)刻，電

56、流 iLm2為零并且二極管Db是斷開(kāi)的。電流ID1 =npiLm1。由于輸出功率等于來(lái)自于電感Lm1和電感Lm2的總和，下面的關(guān)系式為：</p><p>　　從（6），（13）和（24），電流iLm1的最大值和最小值由下式?jīng)Q定</p><p><b>　　4結(jié)束語(yǔ)</b></p><p>　　單級(jí)單刀開(kāi)關(guān)功率因數(shù)校正交/直流轉(zhuǎn)換器已被提出。對(duì)于