電氣外文文獻(xiàn)翻譯---一種低成本智能式電容位置傳感器

1、<p><b>　　本科畢業(yè)論文</b></p><p><b>　　外文文獻(xiàn)及譯文</b></p><p>　　文獻(xiàn)、資料題目：A Low-Cost, Smart Capacitive</p><p>　　Position Sensor</p><p>　　文獻(xiàn)、資料來源：IEEE TR

2、ANSACTIONS ON </p><p>　　INSTRUMENT AND </p><p>　　MEASUREMENT </p><p>　　文獻(xiàn)、資料發(fā)表（出版）日期：1992.12</p><p>　　院 （部）： 信息與電氣工程學(xué)院</p><p>　　專 業(yè)： 電氣工程與自動(dòng)化</p>

3、<p>　　班 級(jí)： 電本064</p><p><b>　　外文文獻(xiàn)：　</b></p><p>　　A Low-Cost, Smart Capacitive Position Sensor</p><p><b>　　Abstract</b></p><p>　　A new h

4、igh-performance, low-cost, capacitive position-measuring system is described. By using a highly linear oscillator, shielding and a three-signal approach, most of the errors are eliminated. The accuracy amounts to 1 μm ov

5、er a 1 mm range. Since the output of the oscillator can directly be connected to a microcontroller, an A/D converter is not needed.</p><p>　　I. INTRODUCTION</p><p>　　This paper describes a novel

6、 high-performance, low-cost, capacitive displacement measuring system featuring:</p><p>　　1 mm measuring range,</p><p>　　1 μm accuracy,</p><p>　　0.1 s total measuring time.</p>

7、;<p>　　Translated to the capacitive domain, the specifications correspond to:</p><p>　　a possible range of 1 pF; </p><p>　　only 50 fF of this range is used for the displacement transducer

8、;</p><p>　　50 aF absolute capacitance-measuring inaccuracy.</p><p>　　Meijer and Schrier [l] and more recently Van Drecht,Meijer, and De Jong [2] have proposed a displacement-measuring system, us

9、ing a PSD (Position Sensitive Detector) as sensing element. Some disadvantages of using a PSD are the higher costs and the higher power consumption of the PSD and LED (Light-Emitting Diode) as compared to the capacitive

10、sensor elements described in this paper.</p><p>　　The signal processor uses the concepts presented in [2],but is adopted for the use of capacitive elements. By the extensive use of shielding, guarding and sm

11、art A/D conversion,the system is able to combine a high accuracy with a very low cost-price. The transducer produces three-period-modulated signals which can be selected and directly read out by a microcontroller. The mi

12、crocontroller,in return, calculates the displacement and can send this value to a host computer (Fig. 1) or a display or dr</p><p>　　Fig. 1. Block diagram of the system</p><p>　　Fig. 2. Perspect

13、ive and dimensions of the electrode structure</p><p>　?、? THE ELECTRODE STRUCTURE</p><p>　　The basic sensing element consists of two simple electrodes with capacitance Cx, (Fig. 2). The smaller

14、one (E2) is surrounded by a guard electrode. Thanks to the use of the guard electrode, the capacitance Cx between the two electrodes is independent of movements (lateral displacements as well as rotations) parallel to th

15、e electrode surface.The influence of the parasitic capacitances Cp will be eliminated as will be discussed in Section Ⅲ.</p><p>　　According to Heerens [3], the relative deviation in the capacitance Cx betwee

16、n the two electrodes caused by the finite guard electrode size is smaller than:</p><p>　　δ<e-π(x/d) (1)</p><p>　　where x is the width of the guard and d the dis

17、tance between the electrodes. This deviation introduces a nonlinearity.Therefore we require that δ is less than 100 ppm.Also the gap between the small electrode and the surrounding guard causes a deviation:</p>&l

18、t;p>　　δ<e-π(d/s) (2)</p><p>　　with s the width of the gap. This deviation is negligible compared to (l), when the gap width is less than 1/3 of the distance between the e

19、lectrodes.</p><p>　　Another cause of errors originates from a possible finite skew angle α between the two electrodes (Fig. 3). Assuming the following conditions:</p><p>　　the potentials on the

20、small electrode and the guard electrode are equal to 0 V,</p><p>　　the potential on the large electrode is equal to V volt,</p><p>　　the guard electrode is large enough,</p><p>　　it

21、 can be seen that the electric field will be concentric.</p><p>　　Fig. 3. Electrodes with angle α.</p><p>　　To keep the calculations simple, we will assume the electrodes to be infinitely large

22、in one direction. Now the problem is a two-dimensional one that can be solved by using polar-coordinates (r, φ). In this case the electrical field can be described by:</p><p><b>　　(3)</b></p&g

23、t;<p>　　To calculate the charge on the small electrode, we set φ to 0 and integrate over r:</p><p><b>　　(4)</b></p><p>　　with Bl the left border of the small electrode:</p&

24、gt;<p><b>　　(5)</b></p><p>　　and Br the right border:</p><p><b>　　(6)</b></p><p>　　Solving (4) results in:</p><p><b>　　(7)</b>

25、;</p><p>　　For small α's this can be approximated by:</p><p><b>　　(8)</b></p><p>　　It appears to be desirable to choose l smaller than d, so the error will depend on

26、ly on the angle α. In our case, a change in the angle of 0.6°will cause an error less than 100 ppm.</p><p>　　With a proper design the parameters εo and l are constant,and then the capacitance between th

27、e two electrodes will depend only on the distance d between the electrodes.</p><p>　　Ⅲ.ELIMINATION OF PARASITIC CAPACITANCES</p><p>　　Besides the desired sensor capacitance C, there are also man

28、y parasitic capacitances in the actual structure (Fig.2). These capacitances can be modeled as shown in Fig.4. Here Cpl represents the parasitic capacitances from the electrode E1 and Cp2 from the electrode E2 to the gua

29、rd electrodes and the shielding. Parasitic capacitance Cp3 results from imperfect shielding and forms an offset capacitance. When the transducer capacitance Cx is connected to an AC voltage source and the current through

30、</p><p>　　Fig. 4. Elimination of parasitic capacitances</p><p>　　The current is measured by the amplifier with shunt feedback, which has a very low input impedance. To obtain the required linear

31、ity, the unity-gain bandwidth fT of the amplifier has to satisfy the following condition:</p><p><b>　　(9)</b></p><p>　　where T is the period of the input signal.</p><p>

32、　　Since Cp2 consists of cable capacitances and the input capacitance of the op amp, it may indeed be larger than Cf and can not be neglected.</p><p>　　IV. THE CONCEPT OF THE SYSTEM</p><p>　　The

33、system uses the three-signal concept presented in [2], which is based on the following principles. When we measure a capacitor Cx with a linear system, we obtain a value:</p><p><b>　　(10)</b><

34、/p><p>　　where m is the unknown gain and Moff, the unknown offset.By performing the measurement of a reference quantity Cref, in an identical way and by measuring the offset, Moff,by making m = 0, the parameter

35、s m and Moff are eliminated.The final measurement result P is defined as:</p><p><b>　　(11)</b></p><p>　　In our case, for the sensor capacitance C, it holds that:</p><p>

36、<b>　　(12)</b></p><p>　　where Ax is the area of the electrode, do is the initial distance between them, ε is the dielectric constant and △d is the displacement to be measured. For the reference el

37、ectrodes it holds that:</p><p><b>　　(13)</b></p><p>　　with Aref the area and dref the distance. Substitution of (12) and (13) into (10) and then into (11) yields:</p><p>

38、;<b>　　(14)</b></p><p>　　Here, P is a value representing the position while a1 and a0 are unknown, but stable constants. The constant a1 =Aref/Ax is a stable constant provided there is a good mech

39、anical matching between the electrode areas. The constant ao = (Arefd0/(Axdref) will also be a stable constant provided that do and dref are constant. These constants can be determined by a one-time calibration. In many

40、applications this calibration can be omitted; when the displacement sensor is part of a larger system, an ove</p><p>　　V . THE CAPACITANCE-TO-PERIOD CONVERSION</p><p>　　The signals which are pro

41、portional to the capacitor values are converted into a period, using a modified Martin oscillator [4] (Fig. 5j.</p><p>　　When the voltage swing across the capacitor is equal to that across the resistor and t

42、he NAND gates are switched off, this oscillator has a period Toff:</p><p>　　Toff = 4RCoff. (15)</p><p>　　Since the value of the resistor is kept constant, the period

43、 varies only with the capacitor value. Now, by switching on the right NAND port, the capacitance CX can be connected in parallel to Coff. Then the period becomes:</p><p>　　Tx=4R(Coff+Cx)=4RCx+Toff

44、 (16)</p><p>　　The constants R and Toff are eliminated in the way described in Section IV.</p><p>　　In [2] it is shown that the system is immune for most of the nonidealities of the

45、op amp and the comparator, like slewing, limitations of bandwidth and gain, offset voltages,and input bias currents. These nonidealities only cause additive or multiplicative errors which are eliminated by the three-sign

46、al approach.</p><p>　　VI. PERIOD MEASUREMENT WITH A MICROCONTROLLER</p><p>　　Performing period measurement with a microcontroller is an easy task. In our case, an INTEL 87C51FA is used,which has

47、 8 kByte ROM, 256 Byte RAM, and UART for serial communication, and the capability to measure periods with a 333 ns resolution. Even though the counters are 16 b wide, they can easily be extended in the software to 24 b o

48、r more.</p><p>　　The period measurement takes place mostly in the hardware of the microcontroller. Therefore, it is possible to let the CPU of the microcontroller perform other tasks at the same time (Fig. 6

49、). For instance, simultaneously with the measurement of period Tx, period Tref and period Toff,the relative capacitance with respect to Cref is calculated according to (11), and the result is transferred through the UART

50、 to a personal computer.</p><p>　　Fig. 5. Modified Martin oscillator with microcontroller and electrodes.</p><p>　　Fig. 6. Period measurement as background process.</p><p>　　Fig. 7.

51、 Position error as function of the position and estimate of the nonlinearity.</p><p>　　VII. EXPERIMENTAL RESULTS</p><p>　　The sensor is not sensitive to fabrication tolerances of the electrodes.

52、 Therefore in our experimental setup we used simple printed circuit board technology to fabricate the electrodes, which have an effective area of 12 mm × 12 mm. The guard electrode has a width of 15 mm, while the di

53、stance between the electrodes is about 5 mm. When the distance between the electrodes is varied over a 1 mm range, the capacitance changes from 0.25 pF to 0.3 pF.Thanks to the chosen concept, even a simple dual op</p&

54、gt;<p>　　The system was tested in a fully automated setup, using an electrical XY table, the described sensor and a personal computer. To achieve the required measurement accuracy the setup was autozeroed every mi

55、nute. In this way the nonlinearity, long-term stability and repeatability have been found to better than 1 μm over a range of 1 mm (Fig.7). This is comparable to the accuracy and range of the system based on a PSD as des

56、cribed in [2].</p><p>　　As a result of these experiments, it was found that the resolution amounts to approximately 20 aF. This result was achieved by averaging over 256 oscillator periods. A further increas

57、e of the resolution by lengthening the measurement time is not possible due to the l/f noise produced by the first stages in both the integrator and the Comparator.</p><p>　　The absolute accuracy can be deri

58、ved from the position accuracy. Since a 1 mm displacement corresponds to a change in capacitance of 50 fF, the absolute accuracy of 1 μm in the position amounts to an absolute accuracy of 50 aF.</p><p>　　CON

59、CLUSION</p><p>　　A low-cost, high-performance displacement sensor has been presented. The system is implemented with simple electrodes, an inexpensive microcontroller and a linear</p><p>　　capac

60、itance-to-period converter. When the circuitry is provided with an accurate reference capacitor, the circuit can also be used to replace expensive capacity-measuring systems.</p><p>　　REFERENCES</p>&

61、lt;p>　　[1] G. C. M. Meijer and R. Schner, “A linear high-performance PSD</p><p>　　displacement transducer with a microcontroller interfacing,” Sensors</p><p>　　and Actuators, A21-A23, pp. 538

62、-543, 1990.</p><p>　　[2]J. van Drecht, G. C. M. Meijer, and P. C. de Jong, “Concepts for the</p><p>　　design of smart sensors and smart signal processors and their application</p><p&g

63、t;　　to PSD displacement transducers,” Digesr of Technical Papers,</p><p>　　Transducers ’91.</p><p>　　[3]W. C. Heerens, “Application of capacitance techniques in sensor design,”</p><p&

64、gt;　　Phys. E: Sci. Insfrum., vol. 19, pp. 897-906, 1986.</p><p>　　[4]K. Martin, ‘‘A voltage-controlled switched-capacitor relaxation oscillator,”</p><p>　　IEEEJ., vol. SC-16, pp. 412-413, 1981.&

65、lt;/p><p><b>　　中文譯文：</b></p><p>　　一種低成本智能式電容位置傳感器</p><p><b>　　摘要</b></p><p>　　本文描述了一種新的高性能，低成本電容位置測(cè)量系統(tǒng)。通過使用高線性振蕩器，屏蔽和三信號(hào)通道，大部分誤差被消除。其精確度在1毫米范圍內(nèi)達(dá)1微米

66、。由于振蕩器的輸出可直接連接到微控制器，所以無需用A/D轉(zhuǎn)換器。</p><p><b>　?、?導(dǎo)言</b></p><p>　　本文介紹了一種新型高性能，低成本的電容位移測(cè)量系統(tǒng)，特點(diǎn)如下：</p><p><b>　　1毫米測(cè)量范圍</b></p><p><b>　　1微米精確度&

67、lt;/b></p><p>　　0.1 s總測(cè)量時(shí)間</p><p>　　對(duì)應(yīng)到電容域，規(guī)格相當(dāng)于：</p><p>　　1皮法的變化范圍；只有這個(gè)范圍的50fF（fF是法拉乘以10的負(fù)15次方。f是femto的縮寫）用于位移傳感器。</p><p>　　50aF絕對(duì)電容測(cè)量誤差。</p><p>　　梅耶爾和

68、施里爾[1]以及最近的范德雷赫特河，梅耶爾，和德容[2]提出了位移測(cè)量系統(tǒng)，采用一個(gè)PSD（位置敏感探測(cè)器）作為傳感元件。和本文描述的電容傳感器元件相比，使用PSD的缺點(diǎn)是，PSD和LED（發(fā)光二極管）有更高的成本和功率消耗。</p><p>　　使用[2]中所提概念的信號(hào)處理器，被采用到電容元件的使用中。通過廣泛使用屏蔽，智能A / D轉(zhuǎn)換，該系統(tǒng)能夠?qū)⒏呔_度和低成本結(jié)合。換能器產(chǎn)生可以選擇和直接由微控制器讀

69、出的三段調(diào)制信號(hào)。微控制器，相應(yīng)的，計(jì)算位移及發(fā)送此值到主機(jī)電腦（圖1）或顯示或驅(qū)動(dòng)執(zhí)行器。</p><p><b>　　圖1 該系統(tǒng)的框圖</b></p><p>　　圖2 電極結(jié)構(gòu)的尺寸和透視圖</p><p><b>　　Ⅱ.電極結(jié)構(gòu)</b></p><p>　　基本傳感元件包含電容為Cx的兩

70、個(gè)簡(jiǎn)單電極(圖2）。較小的一個(gè)（E2）是由屏蔽電極包圍。由于使用屏蔽電極，兩電極間的電容Cx可平行于電極表面獨(dú)立運(yùn)動(dòng)（橫向平移以及旋轉(zhuǎn)）。寄生電容Cp的影響可被消除，將在第3節(jié)討論。</p><p>　　據(jù)Heerens [3]，由有限屏蔽電極大小造成的兩個(gè)電極之間電容Cx的相對(duì)偏差小于：</p><p>　　δ<e-π(x/d)

71、 (1)</p><p>　　其中x是屏蔽的寬度，d是電極之間的距離。這種偏差引入了非線性。因此，我們規(guī)定δ小于100ppm。此外小電極和周圍屏蔽之間的間距產(chǎn)生一個(gè)偏差：</p><p>　　δ<e-π(d/s) (2)</p><p>　　S是間距的寬度。當(dāng)間距寬度小于電極之間距離的1/3

72、時(shí)，這偏差和（1）相比是微不足道的。</p><p>　　另一個(gè)誤差的原因可能源自兩個(gè)電極之間的有限傾斜角α（圖3）。假設(shè)符合下列條件：</p><p>　　小電極和屏蔽電極上的電勢(shì)等于0V</p><p>　　大型電極電勢(shì)等于V伏</p><p><b>　　屏蔽電極足夠大</b></p><p&g

73、t;　　可以看出，電場(chǎng)將同心。</p><p>　　圖3 傾斜角度α的電極</p><p>　　為了使計(jì)算簡(jiǎn)單，我們將假設(shè)電極在一個(gè)方向無限大。問題就成為一個(gè)二維問題，可以用極坐標(biāo)（Υ，φ）方法解決。在這種情況下，電場(chǎng)可以表述為：</p><p><b>　　(3)</b></p><p>　　為了計(jì)算小電極的損耗，我們

74、設(shè)定φ為0，整定Υ：</p><p><b>　　(4)</b></p><p>　　Bl是小電極的左側(cè)邊界：</p><p><b>　　(5)</b></p><p><b>　　Br是右邊界：</b></p><p><b>　　(6)&

75、lt;/b></p><p><b>　　求解（4）結(jié)果：</b></p><p><b>　　(7)</b></p><p><b>　　對(duì)小α的近似：</b></p><p><b>　　(8)</b></p><p>　

76、　選擇比d小的l似乎是可行的，因此該誤差將只決定于角度α。在這種情況下，0.6°的角度變化，將產(chǎn)生小于100 ppm的誤差。</p><p>　　對(duì)參數(shù)εo和l是常數(shù)的設(shè)計(jì)，兩個(gè)電極之間的電容將僅僅取決于電極之間的距離d。</p><p><b>　?、?寄生電容的消除</b></p><p>　　除了理想傳感器電容Cx，在實(shí)際結(jié)構(gòu)中

77、還有許多寄生電容（圖2）。這些電容可以建模，如圖4所示。這里Cpl代表電極El的寄生電容，Cp2是從電極E2到屏蔽電極和屏蔽層的。寄生電容Cp3造成不完善屏蔽，形成一個(gè)偏移電容。當(dāng)傳感器電容Cx連接到AC電壓源，通過電極的電流可測(cè)，Cpl和Cp2，將被消除。Cp3可通過偏移測(cè)量消除。</p><p><b>　　圖4 消除寄生電容</b></p><p>　　電流通過

78、并聯(lián)反饋放大器測(cè)量，它具有非常低的輸入阻抗。要獲取所需的線性度，放大器的單位增益帶寬fT必須符合下列條件：</p><p><b>　　(9)</b></p><p>　　T是在此期間的輸入信號(hào)。</p><p>　　由于Cp2包括電纜電容和運(yùn)算放大器的輸入電容，它很可能大于Cf而不可忽略。</p><p><b&

79、gt;　　Ⅳ.本系統(tǒng)的概念</b></p><p>　　該系統(tǒng)采用了[2]提出的三信號(hào)的概念，它是基于以下原則。當(dāng)我們用線性系統(tǒng)測(cè)量電容Cx，得到一個(gè)值：</p><p><b>　　(10)</b></p><p>　　其中m是未知的增益,Moff是未知偏移。以相同的方式，通過測(cè)量參考量Cref，測(cè)量偏移Moff，使m= 0,參數(shù)

80、m和Moff被抵消。最后的測(cè)量結(jié)果P定義為：</p><p><b>　　(11)</b></p><p>　　在我們的例子中，傳感器的電容Cx為：</p><p><b>　　(12)</b></p><p>　　其中Ax，是電極面積，do是它們之間最初的距離，ε是介電常數(shù)，△d是要測(cè)量的位移。對(duì)

81、于參考電極，它為：</p><p><b>　　(13)</b></p><p>　　Aref為面積，dref為距離。將（12）（13）式代入（10）式，然后代入（11）得：</p><p><b>　　(14)</b></p><p>　　式中，當(dāng)a1和a0未知時(shí)，P是一個(gè)表示位置的值，但是穩(wěn)定的

82、常數(shù)。常數(shù)a1=Aref /Ax是一個(gè)穩(wěn)定的常數(shù)，表明電極之間的區(qū)域有良好的機(jī)械匹配。常量a0=（Arefdo /（Axdref）也是一個(gè)穩(wěn)定的常數(shù)，表明do和dref是常數(shù)。這些常量可以由一次性校準(zhǔn)確定。在許多應(yīng)用中校準(zhǔn)可以省略；當(dāng)位移傳感器是一個(gè)較大系統(tǒng)的一部分，全面的校準(zhǔn)是必須的。這個(gè)整體校準(zhǔn)無需單獨(dú)測(cè)定a1和a0。</p><p>　?、?電容到周期的轉(zhuǎn)換</p><p>　　通過

83、使用改進(jìn)的馬丁振蕩器[4]（圖5），這些與電容值成正比的信號(hào)被轉(zhuǎn)化到一個(gè)周期。</p><p>　　當(dāng)電容兩端的電壓擺幅等于電阻兩端的，NAND門為OFF，這振蕩器有一個(gè)周期Toff：</p><p>　　Toff= 4RCoff. (15)</p><p>　　因?yàn)殡娮柚当３植蛔?，這一周期變化只與電容值有關(guān)?，F(xiàn)

84、在，通過切換在右邊的NAND端口，電容Cx可以平行連接到Coff。即周期為：</p><p>　　Tx=4R（Coff + Cx）= 4RCx+ Toff (16)</p><p>　　常數(shù)R和Toff以在第四節(jié)中描述的方式抵消。</p><p>　　[2]表明，這一系統(tǒng)能克服運(yùn)算放大器和比較器的大多數(shù)非理想狀態(tài)，如回轉(zhuǎn)，限制帶寬

85、和增益，偏移電壓，和輸入偏置電流。這些非理想只造成能被三信號(hào)通道消除的累計(jì)誤差或乘積誤差。</p><p>　?、?用微控制器測(cè)量周期</p><p>　　用微控制器執(zhí)行周期測(cè)量是一件容易的事。在我們的例子中，使用英特爾87C51FA，其中有8kB的ROM，256字節(jié)RAM和串行通信的UART，有333 ns的分辨率進(jìn)行周期測(cè)量。即使計(jì)數(shù)器有16b寬，他們也可以很容易地將軟件擴(kuò)展到24 b

86、或更多。</p><p>　　周期測(cè)量大多發(fā)生在微控制器的硬件中。因此，讓微控制器的CPU在同一時(shí)間執(zhí)行其他任務(wù)是有可能的（圖6）。例如，同時(shí)測(cè)量周期Tx，周期Tref，和周期Toff，關(guān)于Cref相對(duì)電容的計(jì)算方法根據(jù)式（11），其結(jié)果是通過UART轉(zhuǎn)移到個(gè)人計(jì)算機(jī)。</p><p>　　圖5 含微控制器和電極的改進(jìn)馬丁振蕩器</p><p>　　圖6 周期測(cè)量作

87、為背景進(jìn)程</p><p>　　圖7 位移誤差作為位移的函數(shù)及非線性的估計(jì)值</p><p><b>　?、?實(shí)驗(yàn)結(jié)果</b></p><p>　　該傳感器對(duì)電極的制造容差不敏感。因此，在我們的實(shí)驗(yàn)裝置中用簡(jiǎn)單的印刷電路板技術(shù)制造電極，它有一個(gè)12毫米×12毫米的有效區(qū)域。屏蔽電極15毫米寬，電極之間的距離約5毫米。當(dāng)電極之間的距離變

88、化超過1毫米范圍，電容值從0.25 pF變化為0.3pF。所選擇的概念下，甚至可用一個(gè)簡(jiǎn)單的雙運(yùn)放（TLC272AC）和CMOS NAND，允許單5伏電源電壓?？偟臏y(cè)量時(shí)間只有100毫秒，振蕩器運(yùn)行在10千赫下。</p><p>　　該系統(tǒng)用全自動(dòng)化的裝置測(cè)試，使用電力XY工作臺(tái)、描述的傳感器和個(gè)人計(jì)算機(jī)。為了達(dá)到所需的測(cè)量精度，設(shè)置為每分鐘自動(dòng)清零。在這種方式下，非線性、長(zhǎng)期穩(wěn)定和可重復(fù)性，在1毫米范圍內(nèi)精確了

89、1μm（圖7）。這是與[2]所述基于PSD的系統(tǒng)的精確度和范圍相比。</p><p>　　由這些實(shí)驗(yàn)結(jié)果表明，分辨率達(dá)約20aF。這一結(jié)果通過平均超過256個(gè)振蕩周期實(shí)現(xiàn)。通過延長(zhǎng)測(cè)量時(shí)間進(jìn)一步提高分辨率是不可能的，因?yàn)?/ f噪聲產(chǎn)生在積分和比較的第一階段。</p><p>　　絕對(duì)精度決定于定位精度。由于1毫米位移對(duì)應(yīng)電容變化50fF，位移的1μm的絕對(duì)精度數(shù)值即為50aF的絕對(duì)精度。