外文翻譯----基于tms320f2812 dsp的有死區(qū)時(shí)間補(bǔ)償?shù)膕vpwm調(diào)速永磁同步電動(dòng)機(jī)

1、<p>　　Dead-time Compensation of SVPWM Based on DSP TMS320F2812 for PMSM</p><p>　　Song Xuelei*, Wen Xuhui, Guo Xinhua, and Zhao Feng</p><p>　　Institute of Electrical Engineering, Chinese Aca

2、demy of Sciences, Beijing, P.R.China</p><p>　　E-mail: songxl@mail.iee.ac.cn</p><p>　　Abstract—The dead-time effect in a three-phase voltage source inverter can result in voltage losses, current

3、waveform distortion and torque pulsation. In order to improve the current waveform and decrease the torque pulsation, this paper proposes a dead-time compensation method of SVPWM. This method divides the iα - iβ plane in

4、to six sectors and compensates the dead-time of SVPWM according to the sector number of stator current vector determined by the α- and β-axis components of the stator curr</p><p>　　Index Terms--dead-time com

5、pensation,SVPWM,PMSM,TMS320F2812</p><p>　　I. INTRODUCTION</p><p>　　Because the permanent magnet synchronous machine (PMSM) has a lot of advantages such as high power density, high efficiency, hi

6、gh torque to inertia ratio, high reliability, et al[1],therefore, the PMSM driving system have been widely used in many application fields, especially in hybrid electric vehicles (HEV) in recentyears[2]-[6]. In the PMSM

7、driving system, the three-phase voltage source inverter is usually adopted and the IGBT and MOSFET are also used because of their fast switchingfrequency</p><p>　　SVPWM (Space Vector Pulse Width Modulation)

8、is a popular modulation method for three-phase voltage source inverter in motor driving system. In order to improve the current waveform of motors and decrease the torque pulsation of motors, several dead-time compensati

9、on methods of SVPWM have been researched and used in the motor driving system[7]-[11]. Most of the compensation methods are based on the theory of average voltage deviation. In this paper, a novel dead-time compensation

10、method of SVPWM,</p><p>　　II. DEAD-TIME COMPENSATION METHOD</p><p>　　Fig.1 shows the topology diagram of the PMSM driving system whose invert unit adopts the three-phase voltage source inverter.

11、 In Fig.1, Q1, Q2, Q3, Q4, Q5 and Q6 are six IGBTs of the three-phase voltage source inverter, and D1, D2, D3, D4, D5 and D6 are their reverse parallel diodes respectively. In addition, the driving switch signals g1, g2,

12、 g3, g4, g5 and g6 are provided by the control unit of the driving system.</p><p>　　Define the phase currents ia, ib and ic are positive when they flow from the inverter to PMSM, and negative when they flow

13、from PMSM to the inverter. There are eight switch combination states for the six IGBTs in the threephase voltage source inverter, and during the duration of dead-time, there are correspondingly six current combination st

14、ates for three-phase currents ia, ib and ic according to their polarity:</p><p>　　(1) ia>0, ib<0 and ic<0;</p><p>　　(2) ia>0, ib>0 and ic<0;</p><p>　　(3) ia<0,

15、ib>0 and ic<0;</p><p>　　(4) ia<0, ib>0 and ic>0;</p><p>　　(5) ia<0, ib<0 and ic>0;</p><p>　　(6) ia>0, ib<0 and ic>0.</p><p>　　It is ver

16、y important and difficult to detect the zerocross point or the polarity of each phase current.Traditionally, if the zero-cross point is detect directly through A/D converter of DSP or MCU, bigger measurement deviation wi

17、ll be led especially under the condition of small current, which will result in bigger dead-time compensation deviation and also affect the result of dead-time compensation. Therefore, this paper adopts an indirectly met

18、hod to detect the zero-cross point of phase current</p><p>　　For convenient analysis and illustration, place the three-phase currents ia, ib, ic in the three-phase static reference frame and the two current

19、components iα , iβ of the current vector in the two-phase static reference frame into the same figure, which is shown in Fig.2. According to the polarity of three-phase currents ia, ib, ic, the iα - iβ plane in the two-p

20、hase static reference frame can be divided into six sectors: I(1), II(2), III(3), IV(4), V(5) and VI(6). </p><p>　　For each sector in the iα-iβ plane, there is a corresponding dead-time compensation rule. In

21、 other words, once the sector which the current vector belongs to is known, the dead-time effect can be compensated according to the corresponding compensation rule.Therefore, recognizing the sector number of the current

22、 vector is the key problem. </p><p>　　In this paper, the sector number is determined by thecurrent vector angle φ which can be calculated through the α - and β -axis components of the stator current vector.

23、Equation (1) shows the calculation method of the current vector φ, and equation (2) shows the relationship between the sector number and the current vector φ. </p><p>　　φ=kπ+arctan（iβ/iα） (k = 0,1)</p>

24、<p>　　Fig.2. Current Polarity and Current Vector Angle ?</p><p><b>　　TABLE I</b></p><p>　　DEAD-TIME COMPENSATION RULES TABLE OF SVPWM</p><p><b>　　(2)</b&

25、gt;</p><p>　　For three-phase voltage source inverter, the essence of dead-time compensation is to compensating the voltage deviation. However, in the digital motor driving and control system, voltage regulat

26、ion is implemented through pulse width modulation, that is, through regulating the duty cycle of voltage pulse which has something to do with the pulse width T in one PWM period Tpwm. Therefore, in fact it is the pulse w

27、idth T that is compensated in the practical application. TABLE I shows the dead-time c</p><p>　　In one word, the proposed dead-time compensation method can be carried out through the following steps:</p&g

28、t;<p>　　(1) Calculate the current vector angle φ through the α- and β -axis components of the stator current vector in the two-phase static reference frame according to equation (1).</p><p>　　(2) Dete

29、rmine the sector number through the current vector angle φ according to equation (2).</p><p>　　(3) Execute the dead-time compensation algorithm according to the compensation rules table TABLE I.</p>&

30、lt;p>　　III. EXPERIMENTS</p><p>　　In order to test and verify the proposed dead-time compensation method of SVPWM, experiments are established and made. The experiment system consists of PMSM, three-phase

31、voltage source inverter, control platform, dynamometer, heat dissipation system, et al.The type of IGBT in the inverter is CM600DY-24A produced by Mitsubishi. The control platform is based on DSP TMS320F2812 produced by

32、Texas Instrument.It is a special motor control DSP which has many advantages and can implement high-performan</p><p>　　For different pulse width compensation values of 0.76μs,1.10μs,1.33μs and 1.60μs, the de

33、ad-time compensation experiments are all made. Fig.3 shows the experiment waveforms of three-phase stator currents and the sector number of stator current vector for different pulse width compensation values, and Fig.4 s

34、hows the corresponding frequency spectrums.</p><p><b>　　TABLE II</b></p><p>　　MAIN PARAMETERS OF PMSM USED IN EXPERIMENTS</p><p>　　(a) No Compensation</p><p&g

35、t;　　(b) Pulse Width Compensation Value = 0.76μs</p><p>　　(c) Pulse Width Compensation Value =1.10μs</p><p>　　(d) Pulse Width Compensation Value =1.33μs</p><p>　　(e) Pulse Width Comp

36、ensation Value =1.60μs</p><p>　　Fig.3. Experiment Waveforms of Three-phase Stator Currents</p><p>　　Here, the CPU frequency of DSP is set at 150MHz,the switching frequency of IGBTs in three-phas

37、e voltage inverter is set at 10kHz, the dead-time is set at 3.2μs through the hardware and software of DSP,the motor control method adopts FOC algorithm, the dc link voltage is set at about 330V,and the phase current is

38、controlled at about 10 A.</p><p>　　(a) No Compensation</p><p>　　(b) Pulse Width Compensation Value =0.76μs</p><p>　　(c) Pulse Width Compensation Value =1.10μs</p><p>　　

39、(d) Pulse Width Compensation Value =1.33μs</p><p>　　(e) Pulse Width Compensation Value =1.60μs</p><p>　　Fig.4. Frequency Spectrum of Stator Current (Phase A)</p><p>　　It can be seen

40、 from Fig.3 and Fig.4 that, compared with experiment results of no compensation, through the proposed dead-time compensation algorithm the threephase stator current waveforms of PMSM are all improved effectively and the

41、harmonic components of three-phase stator currents are also decreased effectively. Especially when the pulse width compensation value is set at about 1.10μs,compared with experiment results at the other pulse width compe

42、nsation values of 0.76μ,1.33μs and 1.60μs, the</p><p>　　IV. CONCLUSIONS</p><p>　　The proposed dead-time compensation method can be implemented easily through software algorithm without any extra

43、 hardware design. So long as the current vector angle φ is determined by the α- and β-axis components of stator current vector in the two-phase static reference frame, the dead-time compensation algorithm can be carried

44、out effectively according to the corresponding dead-time compensation rules table. Finally experiments are established and made on the PMSM driving platform based on D</p><p>　　REFERENCES</p><p>

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55、es”, IEEE Transactions on Power Electronics, vol.20, No.5, pp. 1150-1160, 2005.</p><p><b>　　外文資料譯文</b></p><p>　　基于TMS320F2812 DSP的有死區(qū)時(shí)間補(bǔ)償?shù)腟VPWM調(diào)速永磁同步電動(dòng)機(jī)</p><p>　　宋雪蕾*溫徐匯

56、，郭新華和趙峰</p><p>　　北京電機(jī)工程學(xué)會(huì)，中國(guó)科學(xué)院，E - mail：songxl@mail.iee.ac.cn</p><p>　　抽象的死區(qū)時(shí)間的影響可導(dǎo)致逆變器三相電壓源電壓損失，電流波形畸變和轉(zhuǎn)矩脈動(dòng)。為了改善目前的波形，并減少轉(zhuǎn)矩脈動(dòng)，提出了一種SVPWM的死區(qū)時(shí)間補(bǔ)償方法。這種方法劃分iα –iβ平面為六個(gè)扇形區(qū)域，進(jìn)入和補(bǔ)償?shù)乃绤^(qū)時(shí)間的SVPWM矢量根據(jù)定子柯部

57、門(mén)數(shù)目這種方法劃分 iα - iβ 平面為6個(gè)扇形區(qū)域，補(bǔ)償?shù)腟VPWM死區(qū)時(shí)間，根據(jù)部門(mén)的α數(shù)和組件的定子β-軸由定子電流矢量的決定。此外，這一方法可以通過(guò)軟件實(shí)現(xiàn)完全沒(méi)有任何額外的硬件數(shù)字信號(hào)處理器TMS320F2812的基礎(chǔ)上最后的實(shí)驗(yàn)，建立和作出的，而實(shí)驗(yàn)結(jié)果表明，該方法是正確和可行的。</p><p>　　關(guān)鍵詞：指數(shù)條款 - 死區(qū)補(bǔ)償，SVPWM的，永磁同步電機(jī)，TMS320F2812</p&

58、gt;<p><b>　　引言</b></p><p>　　由于永磁同步電機(jī)（PMSM的）有很多優(yōu)勢(shì)，例如，高功率密度，高效率，高慣性力矩比，高可靠性等[1]，因此，永磁同步電機(jī)驅(qū)動(dòng)系統(tǒng)已被廣泛應(yīng)用于許多應(yīng)用領(lǐng)域，尤其是在最近幾年應(yīng)用在混合動(dòng)力（HEV）用電動(dòng)汽車(chē)上[2] - [6]。</p><p>　　在永磁同步電機(jī)驅(qū)動(dòng)系統(tǒng)，三相電壓源逆變器通常采用

59、的IGBT和MOSFET也因?yàn)樗鼈兊拈_(kāi)關(guān)頻率而普遍使用。為了避免短路的同時(shí)轉(zhuǎn)向裝置的直流環(huán)節(jié)發(fā)生的時(shí)候，雙方在同一階段的切換，死區(qū)時(shí)間通常是在門(mén)信號(hào)驅(qū)動(dòng)開(kāi)關(guān)的時(shí)候。在持續(xù)死區(qū)時(shí)間中，相都相同的兩個(gè)開(kāi)關(guān)裝置處于關(guān)閉狀態(tài)。當(dāng)時(shí)現(xiàn)有的死將導(dǎo)致一系列問(wèn)題的死區(qū)時(shí)間的影響，例如，電壓損失，電流波形畸變和轉(zhuǎn)矩脈動(dòng)，特別是在高速條件下的小電流或低。</p><p>　　空間矢量脈寬調(diào)制（空間矢量脈寬調(diào)制）在電機(jī)逆變器是一種流行

60、的調(diào)制方式為3相電壓源驅(qū)動(dòng)系統(tǒng)。為了提高電動(dòng)機(jī)的電流波形，降低電機(jī)轉(zhuǎn)矩脈動(dòng)，幾個(gè)死區(qū)時(shí)間補(bǔ)償?shù)腟VPWM方法進(jìn)行了研究和系統(tǒng)在駕駛汽車(chē)[7]-[11]. 大部分的補(bǔ)償辦法是根據(jù)偏差理論的平均電壓。</p><p>　　在此提出了一種新穎的死區(qū)時(shí)間補(bǔ)償?shù)腟VPWM方法，這也是基于平均電壓的偏差理論。這種方法劃分iα - iβ平面成6個(gè)部分，并彌補(bǔ)了時(shí)間的SVPWM根據(jù)定子電流矢量根據(jù)定子電流部門(mén)角度φ。確定的α -

61、和β -定子軸的α組成部分的電流矢量中- β參照系。另外，該方法通過(guò)軟件可以實(shí)現(xiàn)完全沒(méi)有任何額外的硬件設(shè)計(jì)。最后的實(shí)驗(yàn)，是基于數(shù)字信號(hào)處理器TMS320F2812的駕駛平臺(tái)，測(cè)試驗(yàn)證了提出的PMSM和補(bǔ)償方法。</p><p><b>　　死區(qū)補(bǔ)償方法</b></p><p>　　圖1顯示了逆變器的拓?fù)鋱D的永磁同步電機(jī)驅(qū)動(dòng)系統(tǒng)的轉(zhuǎn)化裝置采用了三相電壓。在圖1，Q1，Q

62、2，Q3，Q4，Q5和Q6有6逆變器的IGBT的三相電壓源，和D1，D2和D3，D4，D5和D6中的反向平行二極管。另外, 開(kāi)關(guān)的驅(qū)動(dòng)信號(hào)G1，G2，G3，G4，G5和G6是系統(tǒng)提供的驅(qū)動(dòng)控制裝置。</p><p>　　定義相電流Ia，Ib和Ic從永磁同步電動(dòng)機(jī)變頻流向?yàn)檎?，?dāng)決定逆變流流向永磁同步電動(dòng)機(jī)為負(fù)時(shí)。有8個(gè)開(kāi)關(guān)的三相電壓源逆變器的六個(gè)IGBT組合狀態(tài)，并在死區(qū)時(shí)間中，有相應(yīng)的6個(gè)當(dāng)前結(jié)合態(tài)對(duì)應(yīng)三相電流

63、IA，IB和IC根據(jù)自己的極性：</p><p>　　(1) ia >0, ib <0 and ic <0; </p><p>　　(2) ia >0, ib >0 and ic <0; </p><p>　　(3) ia <0, ib >0 and ic <0;</p><p>　　(4

64、) ia <0, ib >0 and ic >0;</p><p>　　(5) ia <0, ib <0 and ic >0;</p><p>　　(6) ia >0, ib <0 and ic >0.</p><p>　　圖1。拓?fù)鋱D的永磁同步電機(jī)驅(qū)動(dòng)系統(tǒng)</p><p>　　零交叉點(diǎn)或

65、每個(gè)階段極性電流是非常重要和難以檢測(cè)的。照慣例，如果零交叉點(diǎn)檢測(cè)單片機(jī)直接通過(guò)數(shù)字信號(hào)處理器或/ D轉(zhuǎn)換器，較大的測(cè)量誤差將導(dǎo)致特別是在小電流條件下，這將導(dǎo)致更大的死區(qū)時(shí)間補(bǔ)償?shù)钠?，也影響了死區(qū)時(shí)間補(bǔ)償結(jié)果。因此，本文采用一種間接的方法來(lái)檢測(cè)零交叉點(diǎn)，是在兩相靜止坐標(biāo)系上基于電流矢量角φ來(lái)檢測(cè)的。</p><p>　　為方便分析和說(shuō)明，定義三相電流ia,ib,ic在三相靜止坐標(biāo)系的兩個(gè)電流分量iα,iβ在兩相靜

66、止坐標(biāo)系電流矢量有相同的數(shù)字，這顯示在圖2。根據(jù)三相電流ia,ib,ic的極性, 兩相靜止坐標(biāo)系iα-iβ可分為6個(gè)部分：I(1),II(2),III(3),IV(4),V(5) 和VI(6).</p><p>　　對(duì)于兩相靜止坐標(biāo)系iα-iβ每個(gè)部分，有相應(yīng)的死區(qū)時(shí)間補(bǔ)償規(guī)則。換句話說(shuō)，一旦該部分的電流矢量屬于已知，死區(qū)時(shí)間可以根據(jù)相應(yīng)的補(bǔ)償規(guī)則得到補(bǔ)償。因此，認(rèn)識(shí)到當(dāng)前的矢量扇區(qū)數(shù)是關(guān)鍵問(wèn)題。</p&g

67、t;<p>　　本文，該扇形的數(shù)目取決于電流矢量角φ，它可以通過(guò)計(jì)算 α - 和β-軸定子組件的電流矢量來(lái)得到。方程（1）顯示當(dāng)前向量φ的計(jì)算方法，和方程（2）顯示了扇形和電流矢量φ之間的數(shù)量關(guān)系。</p><p>　　φ=kπ+arctan（iβ/iα） (k = 0,1)</p><p>　　圖2 電流極性和電流矢量角φ</p><p>　　表一

68、 SVPWM死區(qū)時(shí)間補(bǔ)償規(guī)則表</p><p>　　對(duì)于三相電壓源逆變器，對(duì)死區(qū)時(shí)間的本質(zhì)補(bǔ)償，是補(bǔ)償電壓偏差。然而，在數(shù)字電機(jī)驅(qū)動(dòng)和控制系統(tǒng)中，電壓調(diào)節(jié)是通過(guò)脈沖寬度調(diào)制，即通過(guò)調(diào)節(jié)占空比脈沖電壓，它是與脈沖寬T在一個(gè)脈寬調(diào)制的周期Tpwm。因此，事實(shí)上它是脈沖寬度T的實(shí)際應(yīng)用中得到補(bǔ)償。表格一顯示死區(qū)時(shí)間補(bǔ)償規(guī)則相應(yīng)的三相電流極性ia,ib,ic和扇區(qū)數(shù)目前的矢量在iα-iβ平面?？梢钥闯觯瑢?duì)于iα-iβ坐標(biāo)

69、系的不同扇區(qū)，相應(yīng)的補(bǔ)償值是不同的。</p><p>　　一句話，建議的死區(qū)時(shí)間補(bǔ)償方法可以進(jìn)行通過(guò)以下步驟：</p><p>　　在兩相靜止坐標(biāo)系根據(jù)方程（1）通過(guò)α-和β-軸組件計(jì)算定子電流矢量的電流矢量角φ。</p><p>　　通過(guò)電流矢量角φ根據(jù)方程（2）確定的部門(mén)數(shù)目</p><p>　　根據(jù)表一執(zhí)行的死區(qū)補(bǔ)償算法的補(bǔ)償規(guī)則。&l

70、t;/p><p><b>　　三．實(shí)驗(yàn)</b></p><p>　　為了測(cè)試和驗(yàn)證所提出的停滯時(shí)間補(bǔ)償?shù)腟VPWM方法，實(shí)驗(yàn)建立了。該實(shí)驗(yàn)系統(tǒng)由永磁同步電動(dòng)機(jī)，三相電壓源逆變器，控制平臺(tái)，功率計(jì)，散熱系統(tǒng)等組成。IGBT逆變器類(lèi)型是CM600DY - 24A，三菱公司生產(chǎn)?？刂破脚_(tái)是基于DSP TMS320F2812的，德州儀器生產(chǎn)。這是一個(gè)特殊的DSP控制的電機(jī)，具有許

71、多優(yōu)點(diǎn)，并能實(shí)現(xiàn)高性能的電機(jī)控制，例如磁場(chǎng)定向控制（磁場(chǎng)定向控制）和DTC（直接轉(zhuǎn)矩控制）。對(duì)控制對(duì)象永磁同步電動(dòng)機(jī)用于實(shí)驗(yàn)的主要參數(shù)列于表二。</p><p>　　對(duì)于不同的脈沖寬度補(bǔ)償值0.76μs , 1.10μs , 1.33μs以及1.60 μs的死區(qū)補(bǔ)償?shù)膶?shí)驗(yàn)都完成了。圖3顯示了三相定子電流實(shí)驗(yàn)波形和不同的脈沖寬度補(bǔ)償值對(duì)定子電流矢量部門(mén)數(shù)的影響，圖4顯示了相應(yīng)的頻譜。</p><

72、p>　　表二 永磁同步電動(dòng)機(jī)主要技術(shù)參數(shù)和實(shí)驗(yàn)</p><p><b>　　(a) 無(wú)補(bǔ)償</b></p><p>　　(b) 脈沖寬度補(bǔ)償值= 0.76μ的</p><p>　　脈沖寬度補(bǔ)償值= 1.10μ的</p><p>　　脈沖寬度補(bǔ)償值= 1.33μ的</p><p>　　脈沖

73、寬度補(bǔ)償值= 1.60μ的</p><p>　　圖3。實(shí)驗(yàn)波形三相定子電流</p><p>　　在這里，DSP的CPU的頻率為150MHz，IGBT的開(kāi)關(guān)的三相電壓型逆變器頻率為10kHz，通過(guò)硬件和DSP軟件死區(qū)時(shí)間定為3.2μs，F(xiàn)OC電機(jī)控制方法采用的算法，鏈接的直流電壓設(shè)置為約330V，和相電流控制在10A。</p><p><b>　　無(wú)補(bǔ)償&l

74、t;/b></p><p>　　脈沖寬度補(bǔ)償值= 0.76μ的</p><p>　　脈沖寬度補(bǔ)償值= 1.10μ的</p><p>　　脈沖寬度補(bǔ)償值= 1.33μ的</p><p>　　脈沖寬度補(bǔ)償值= 1.60μ的</p><p>　　圖4。定子電流的頻率（相位譜一）</p><p>

75、　　從圖3和圖4可以看出，與不予補(bǔ)償試驗(yàn)結(jié)果相比較，通過(guò)擬議的死區(qū)時(shí)間補(bǔ)償算法的三相永磁同步電動(dòng)機(jī)定子電流波形都有效地提高，三相定子電流的諧波成分，也有效地降低。尤其是當(dāng)脈沖寬度補(bǔ)償價(jià)值被設(shè)置約1.10μs時(shí)，相比于其他補(bǔ)償值的寬度脈沖的實(shí)驗(yàn)結(jié)果如0.76μs，1.33μs以及1.60 μs，補(bǔ)償結(jié)果是最好的，而且三相定子電流的諧波成分是最少的。因此，建議死區(qū)補(bǔ)償?shù)姆椒ㄊ钦_和可行的。</p><p><b

76、>　　四，總結(jié)</b></p><p>　　擬議的死區(qū)時(shí)間補(bǔ)償方法可以通過(guò)軟件算法很容易實(shí)現(xiàn)無(wú)需任何額外硬件設(shè)計(jì)。只要目前的矢量角φ是由定子電流矢量在兩相靜止坐標(biāo)系上的α- 和β-軸組件決定，死區(qū)補(bǔ)償算法可以按相應(yīng)停滯時(shí)間有效地進(jìn)行。最后的實(shí)驗(yàn)是建立完善了基于DSP TMS320F2812的驅(qū)動(dòng)永磁同步電動(dòng)機(jī)作平臺(tái)，其結(jié)果表明，該方法可以改善目前的失真，降低扭矩，尤其是當(dāng)脈沖寬度補(bǔ)償值等于約1.

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